In the optical communications space, techniques used to detect data modulated onto an optical signal may be broadly grouped into two classes, namely “direct” detection and “coherent” detection. In “direct” detection techniques, the optical signal is made incident on a photodetector. The electrical current appearing at the photodetector output is proportional to the optical power. Data modulated onto the optical signal power using an amplitude-modulation scheme, such as On-Off Keying (OOK) can thus be detected by analysis of the photodetector output current. Direct detection techniques have advantages in terms of low cost, and high reliability for On-Off Keying (OOK) based modulation schemes. As a result, the majority of optical receivers currently used in optical communications networks are based on direct detection.
In “coherent” detection techniques, the optical signal is mixed with a strong, narrow-line-width, local oscillator signal by an optical hybrid, and the combined signal made incident on one or more photodetectors. In some systems, the inbound optical signal is first split into orthogonal polarizations, and each polarization processed by a respective optical hybrid. In-phase and Quadrature components of each polarization can be detected using respective photodetectors positioned to receive corresponding signals output by the optical hybrid. The frequency spectrum of the electrical current appearing at the photodetector output(s) is substantially proportional to the convolution of the received optical signal and the local oscillator, and contains a signal component lying at an intermediate frequency that contains the data. Consequently, this “data component” can be isolated and detected by electronically filtering and processing the photodetector output current.
Coherent detection receivers offer numerous advantages over direct detection receivers, many of which follow from the fact that coherent detection techniques provide both phase and amplitude information of the optical signal. As such, more robust modulation schemes, such as binary phase shift keying (BPSK), quadrature phase shift keying (QPSK), and quadrature amplitude modulation (QAM) can be used.
However, receivers based on coherent detection techniques have suffered disadvantages that have, to date, prevented successful deployment in “real-world” installed communications networks. In particular, both the transmitted carrier signal and the local oscillator signal are generated by respective Tx and LO lasers, which, in the case of “real world” network systems, will be semi-conductor laser. As is well known in the art, such lasers exhibit a finite line width and non-zero phase noise. Semiconductor lasers typically used in optical communications system are governed by a control loop which maintains a desired average laser output frequency. However, frequency transients as high as ±400 MHz at rates of up to 50 kHz are common. In addition, many such lasers often exhibit a maximum line width tolerance of about ±2 MHz. As a result, even when the Tx and LO lasers are operating at nominally the same frequency, a mismatch or offset of as much as ±4 MHz can still exist. Short period phase noise in both of the Tx and LO lasers may significantly increase the frequency mismatch beyond this amount.
In a coherent receiver, a frequency mismatch between the received carrier (that is, the Tx laser) and the LO appears as a time varying phase error of detected symbols. When the phase error reaches π/4 for QPSK or π/2 for BPSK, a “cycle-slip” can occur, in which symbols can be erroneously interpreted as lying in an adjacent quadrant. This can result in the erroneous interpretation of every symbol (and thus all data) following the cycle-slip. Accordingly, it is desirable to be able to track and compensate the frequency offset between the Tx carrier and LO signal frequencies. A carrier recovery circuit capable of generating the required carrier error signal, even for BPSK/QPSK signals in which the carrier is suppressed, is described in “Phase Noise-Tolerant Synchronous QPSK/BPSK Baseband-Type Intradyne Receiver Concept With Feedforward Carrier Recovery”, R Noé, Journal of Lightwave Technology, Vol. 23, No. 2, February 2005, and illustrated in FIG. 1.
As may be seen in FIG. 1, an optical signal received through an optical link 2 is divided by a polarization beam splitter 4 into orthogonal polarizations (nominally referred to as X and Y polarizations in FIG. 1), which are then mixed with a local oscillator (LO) 6 through a quadrature 90° optical hybrid 8. The composite optical signals appearing at the output of the hybrid 8 are made incident on a set of photodetectors 10 to generate analog electrical signals Re(X), Im(X), Re(Y), Im(Y) respectively corresponding to real (Re) and imaginary (Im) parts of each polarization. The analog polarization signals X (=Re(X)+jIm(X)) and Y (=Re(Y)+jRe(Y)) are then supplied to a respective analog carrier recovery circuit 12. For simplicity of illustration, only the carrier recovery circuit 12 for the X-polarization is shown in FIG. 1, it being understood that the carrier recovery circuit for the Y-polarization is substantially identical. As may be seen in FIG. 1, the carrier recovery circuit 12 utilizes cascaded frequency doublers 14 (such as Gilbert multiplier cells) which increase the X-polarization frequency by a factor of four. A filter 16 is used to remove broadband noise, and cascaded regenerative frequency dividers 18 used to divide the frequency of the filtered signal by four to obtain a complex recovered carrier signal C*, which is then mixed (at 20) with the complex polarization signal X to obtain the complex baseband signal D.
It is important to note that the X and Y polarizations in Noé are the received polarizations. These do not in general match the transmitted polarizations due to the polarization dynamics of the fiber.
A limitation of this arrangement is that the analog carrier recovery circuit 12 is an inherently narrow-band device. In particular, the Intermediate Frequency (IF) linewidth of the carrier recovery circuit 12 is proportional to the bit error rate (BER): increasing the IF linewidth increases the frequency mis-match that can be corrected by the carrier recovery circuit 12, but at a cost of increasing the BER of the baseband signal D. The moderate-to-severe impairments (e.g. chromatic dispersion, Inter-Symbol Interference-ISI, Polarization Dependent Loss-PDL, Polarization Mode Dispersion-PMD, etc.) encountered in “real-world” installed networks compound this difficulty. Even without an impaired optical signal, Noé's experimental results suggest that in order to achieve an industry-standard 10−9 BER, the maximum permissible IF linewidth would have to be on the order of 1.8 MHz (for a 10 Gbaud QPSK single-polarization system). Such a low IF linewidth necessitates the use of high-precision lasers having a line width of ≦1 MHz, and extremely low transient excursions from the desired frequency. Such lasers are very expensive, and thus are not normally used in communications systems. Clearly, the methods of Noécannot be use with semiconductor lasers of the type commonly used in communications networks, having a line width on the order of ±2 MHz and frequency transients of ±400 MHz.
An additional limitation of Noé's system is that analog circuits are notoriously well known for their inability to adapt to changes following manufacture. At best, the analog carrier recovery circuit of Noé may be able to compensate for performance drift due to component heating, and possibly aging effects. However, a carrier recovery circuit optimized for 10 Gbaud single-polarization signals will not be able to accommodate a polarization-multiplexed 40 GBaud signal.
Accordingly, methods and techniques that enable carrier recovery in a receiver unit of an optical network remain highly desirable.